In the optical communications space, various techniques are used to synthesize an optical communications signal for transmission. A popular technique utilizes a laser 2 coupled to an external optical modulator 4, as shown in FIG. 1a. The laser 2 generates a narrow-band continuous wave (CW) optical carrier signal 6 having a desired wavelength. The optical modulator 4 operates to modulate the amplitude and/or phase the carrier signal 6 to generate the optical communications signal 8 based on a drive signal 10 that encodes data to be transmitted. Typically, the drive signal 10 is generated by a driver circuit 12, which normally provides a power amplifier for amplifying the power of an input digital data signal x(m) to satisfy the input power requirements of the modulator 4.
In the arrangement illustrated in the FIG. 1a, the optical modulator 4 is provided by a well known Mach-Zehnder (MZ) interferometer. Other types of modulators may be used, depending on the desired type of modulation. For example, an electro-absorptive modulator (EAM) may be used for amplitude modulation; whereas phase modulators are well known for implementing phase modulation. In each case, the driver circuit 12 generates the drive signal 10 by scaling the input data signal x(t) to satisfy the voltage and current requirements of the modulator 4. The driver circuit 12 may also generate one or more bias signals (not shown) for controlling a bias point of the modulator 4 in a manner well known in the art.
FIGS. 1b and 1c illustrate a typical eye-opening diagrams of the optical communications signal 8 at the output of the optical modulator 4 and at a receiver end 14 of an optical link 16, respectively. As may be seen in FIG. 1b, at the modulator output, the eye-opening 18 is dominated by the data modulation, with little if any noise, most of which is contributed by laser phase noise and signal leakage through the optical modulator 4. As the optical signal traverses the link 16, the optical signal power is attenuated due to absorption and incomplete amplification. At the same time, signal distortions accumulate due to residual (uncompensated) dispersion and PDL, as well as noise introduced by imperfect optical components (primarily the amplifiers) within the link 16. As a result, the optical signal at the receiver end 14 of the link 16 exhibits a comparatively narrow eye-opening 18′, as a combined result of residual distortions and accumulated noise, as may be seen in FIG. 1c. Comparison between FIGS. 1b and 1c illustrates that the Optical Signal to Noise Ratio (OSNR) at the receiver end 14 of the link 16 is significantly lower than at the transmitter, and this typically holds true even for links with significant optical dispersion compensation and channel equalization devices within the link.
FIG. 2 illustrates an alternative arrangement known, for example, from Applicant's co-pending U.S. patent application Ser. No. 10/677,223 filed Oct. 3, 2003. In that system, a complex driver circuit 20 comprises a digital filter 22 which uses the input data signal x(m) and a compensation function c(t) to calculate multi-bit In-Phase and Quadrature component values I(n) and Q(n) of a target optical E-field modulation. A non-linear compensator 24 uses the I(n) and Q(n) components to compute multi-bit sample streams VR(n) and VL(n). These digital sample streams are then converted into corresponding analog voltage levels by respective multi-bit digital-to-analog converters (DACs) 26, filtered (at 28) to reduce out-of-band noise, and scaled by low noise amplifiers 30 to yield a pair of drive signals VR(t) and VL(t) which are supplied to respective branches of the MZ modulator 4. If desired, respective digital filters (not shown) may be positioned between the non-linear compensator 24 and the DACs 26 in order to compensate any propagation delay differences between the DACs 26 and the MZ modulator 4.
The arrangement of FIG. 2 is particularly advantageous in that the multi-bit sample values VR(n) and VL(n) can be computed taking into account non-linearities of the analog signal path (e.g. the DACs 26, filters 28 and LNAs 30) and the MZ modulator 4, such that the optical E-field of the composite signal 8 appearing at the output of the MZ modulator 4 closely matches the target E-field modulation computed by the digital filter 22. Additionally, the compensation function c(t) can be selected to compensate impairments of the optical link 16, in which case the target E-field modulation represents a pre-distorted signal which will be transformed by the link impairments into a substantially undistorted optical signal at a receiver end 14 of the link 16.
FIG. 2b illustrates the eye-opening of the optical signal at the receiver end 14 of the link 16. As noted above, distortions due to at least dispersion and polarization effects are substantially eliminated, leaving only amplifier (and other imperfect device) noise as the dominant noise contribution. The result is a wide eye-opening 18″ at the receiver end 14 of the link 16, as may be seen in FIG. 2b. In practice, an optical transmitter of the type illustrated in FIG. 2 has been shown to be capable of pre-compensating in excess of 30000 pS/nm of dispersion, as well as polarization dependent losses. In some implementations, the accuracy of pre-compensation is high enough to obtain a raw bit error rate (BER), that is, before Forward Error Correction (FEC), on the order of 10−12 using a conventional direct detection receiver.
Such a low raw BER exceeds the requirements of most optical transmission protocols (e.g., 10−9 for SONET), which, superficially at least, would appear to be a good thing. However, in practice the raw BER is used as a data input for various control processes used in the receiver. If the raw BER is too low, these control processes may drift to a sub-optimal condition due to a lack of input data.
For example, FIG. 3a is a block diagram schematically illustrating a conventional direct detection receiver 14. As shown in FIG. 3a, an optical signal 8′ received through the optical link 16 is made incident of a photodetector 32 to generate a corresponding analog electrical signal 34. The electrical signal 34 is then amplified (at 36) and filtered (at 38) to remove out-of-band noise, and then supplied to a detector circuit 40. In general, the detector circuit 40 operates as an analog-to-digital (A/D) converter, which compares the signal level to a slicing threshold VTh, at a selected sample phase φ, to obtain a corresponding digital signal. A Forward Error Correction (FEC) circuit 42 then processes the digital signal in a known manner to correct errored data bits.
Typically, the slicing threshold VTh and sample phase φ are controlled on the basis of the raw BER detected by the FEC circuit 42. In general, a control loop 44 is implemented which adjusts the slicing threshold VTh and sample phase φ to optimize the BER, which normally results in the slicing threshold VTh and sample phase φ lying in the center of the eye-opening 18, as may be seen in FIG. 3b. However, if the raw BER is too low, the control loop 44 will have insufficient input data to accurately determine the actual sample phase φ of the detector circuit 40, and thus cannot make appropriate decisions as whether (and in which direction) the sample phase should be adjusted. For example, consider the above-noted case where the raw BER at the receiver 14 is on the order of 10−12. At a line rate of 10 Gbps, this translates into one errored bit every 1000 seconds. During that interval, the control loop 44 receives no input data, and thus makes no adjustments to the sample phase φ. As a result, the sample phase φ can drift (undetectably) due to small differences between the data rate and the receiver's reference clock not shown. This normally results in the sample phase φ drifting to one end of the eye-opening, as may be seen in FIG. 3c, where the reduced signal-to-noise-ratio (SNR) raises the BER to a level at which the control loop 44 can function. While the detector 40 and FEC circuit 42 can recover digital data with the sample timing in this condition, the noise margin of the receiver 14 is significantly reduced. As a result, optical signal transients which increase the SNR can result in loss of data.
Accordingly, methods and apparatus enabling a BER-based control loops to function in the presence of an optical signal transmitter capable of reducing BER to extremely low levels remain highly desirable.